1. Field of the Invention
The present invention relates to a current oscillation type switching power circuit having improved power factor, for example.
2. Description of Related Art
Recently, switching elements which are capable of handling to relatively large currents or voltage of high frequency have been developed, and this development of the switching elements promotes the shift of most of power units for rectifying commercial power to obtain a desired DC voltage to switching type power units. The switching power source enables transformers and other equipment to he miniaturized by increasing its switching frequency, and it is used as a large-power DC--DC converter acting as a power source for various electronic equipment.
When commercial power is rectified, current flowing in a smoothing circuit generally has a distorted waveform, and thus there occurs a problem that a power factor indicating the efficiency of the power source is damaged. Furthermore, a countermeasure to suppressing higher harmonics produced due to distorted current waveform is also required. In order to improve the power factor of the power source, it is the simplest manner to use a rectifying circuit having a choke input system, for example, and it is also preferable for the purpose of the countermeasure of electromagnetic noise (EMI). Furthermore, there is also considered a Magnet-Switch system (hereinafter referred to as "MS system" in which an average charging voltage of a smoothing capacitor is reduced by using the interrupted voltage of the switching power source, and a conducting angle of a rectifying element is broadened to improve the power factor.
The applicant of this application has previously proposed an invention in which a power factor improving means of the MS system is applied to a switching power circuit using a current oscillation type converter (Japanese Patent Application No. Hei-6-210740). FIG. 1 is a circuit diagram showing an example of a switching power circuit which is constructed on the basis of the above invention. In this case, it comprises a separately exciting current oscillation converter based on a half bridge.
In FIG. 1, AC represents a commercial alternating power source. An LC low pass filter comprising impedance elements such as a filter choke coil LN and a filter capacitor CN is provided for the alternating power source AC, and it is used to prevent high-frequency noise of a switching frequency from flowing into an AC line. D1 represents a bridge rectifying circuit comprising four diodes, and it performs full-wave rectification on the input alternating power AC. A high-speed recovery type (represented by DFR) is used for each of the two rectifying diodes as indicated by broken lines, and this is because a high-frequency current of the switching frequency as described later flows in a full-wave rectified output line. The full-wave rectified output is charged through a choke coil CH and a tertiary winding N3 into a smoothing capacitor Ci.
Q1, Q2 represents a switching element constituting a half bridge type switching circuit, and in this case it comprises an MOS-FET transistor. These switching elements are connected to each other in series between the positive side of the smoothing capacitor Ci and ground. The switching elements Q1 and Q2 are driven to perform such a switching operation as to be alternately switched on and off by an oscillation drive circuit 2. Each of DD1 and DD2 which are provided in parallel to the switching elements Q1 and Q respectively represents a clamp diode which forms a current path at a switch-off time. Reference numeral 3 represents an actuating circuit. The actuating circuit starts its operation, for example, when the charging for the smoothing capacitor Ci is started at a power-on time and a charge voltage appears in the smoothing capacitor Ci, and actuates the oscillation drive circuit 2.
PIT represents an insulating transformer for transmitting the switching outputs of the switching elements Q1 and Q2 to a secondary side. One end of the primary winding N1 of the insulating transformer is connected to a connection point of the source/drain of the switching elements Q1 and Q2 through a resonance capacitor C1, and the other end is grounded. A series resonance circuit is formed by an inductance component of the insulating transformer PIT containing the resonance capacitor C1 and the primary winding N1. In this case, the primary winding N1 is wound up to form a winding N4, and the winding N4 is connected to a rectifying and smoothing circuit comprising diodes D4 and the capacitor C4 as shown in FIG. 1. Accordingly, direct current obtained in the rectifying and smoothing circuit is supplied to the actuating circuit 3.
Furthermore, a tertiary winding N3 is wound around the insulating PIT, and a switching voltage V3 which is induced by the tertiary winding N3 is applied across the choke coil CH and the smoothing capacitor Ci. That is, the switching voltage is supplied to a charging path of the smoothing capacitor Ci. Accordingly, after passing through the choke coil CH, the rectified full-wave rectified voltage is superposed on the switching voltage and then charged into the smoothing capacitor Ci.
At the secondary side of the insulating PIT, the induced voltage of the secondary winding N2 which is based on the primary winding N1 is converted to a DC voltage by the bridge rectifying circuit D3 and the smoothing capacitor C3, and it is set as an output voltage Eo. A control circuit 1 compares the DC voltage output Eo of the secondary side with a reference voltage, and supplies the oscillation drive circuit 2 with a control signal corresponding to an error (difference) between the DC voltage output Eo and the reference voltage. In the oscillation drive circuit 2, for example, the switching frequency is varied in accordance with the control signal to thereby perform a constant voltage control operation.
In the switching power circuit thus constructed, since the tertiary winding N3 is provided to the insulating transformer PIT in which the resonance current flows in the primary winding N1, the switching voltage which is excited by the tertiary winding N3 is supplied to the charging path of the smoothing capacitor Ci, and superposed on the rectified voltage. Accordingly, the flow angle of the current flowing out from the bridge rectifying circuit D1 is enlarged, so that the average value thereof becomes a charging current which is close to a sine wave. As a result, the alternating current which is supplied from the commercial alternating power source is reduced in distortion of higher harmonics, and the power factor is improved.
The current 11 which flows out from the rectifying circuit is interrupted at the switching period, and flows discontinuously. Therefore, a high-speed recovery type is required to be used for any two diodes of the bridge rectifying circuit D1. In FIG. 1, two diodes at the anode side as indicated by a broken line DFR are the high-speed recovery type.
Furthermore, the applicant of this application previously proposed a switching power unit in which an magnetically-coupled transformer for exciting a voltage corresponding to a switching output in a choke coil from a primary side or secondary side of an insulating transformer is provided, and the rectified output of the bridge rectifying circuit is superposed on the voltage of the switching period by the magnetically-coupled transformer, thereby improving the power factor (Japanese Patent Application No. Hei-6-192737). With this switching power unit, variation of the rectified and smoothed voltage Vi can be more easily controlled by the circuit shown in FIG. 1.
FIG. 2 is a circuit diagram showing an example of a switching power unit having the magnetically-coupled transformer as described above. In this case, the switching power unit is based on a half bridge type of self-exciting type current resonance switching power unit using transistors for the switching elements Q1 and Q2. The same elements as FIG. 1 are represented by the same reference numerals, and the description thereof is omitted.
The switching elements Q1 and Q2 of this circuit are connected to each other through their collectors and emitters between the connection point of the anode side of the smoothing capacitor Ci and ground. R6 represents an actuating resistor, and DD1 and DD2 which are interposed between each base-emitter of the switching elements Q1 and Q2 respectively represent damper diodes. R5 represents a base current (drive current) adjusting resistor for the switching elements Q1 and Q2. C5 represents resonance capacitor, and it forms a resonance circuit for self-exciting oscillation together with driving windings NB of a drive transformer PRT as described later.
PRT represents a drive transformer for variably controlling the switching frequency of the switching elements Q1 and Q2, and in the case of FIG. 2, it is designed as an orthogonal type saturable reactor in which driving windings Nb and a resonance current detection winding ND are wound around the drive transformer, and further a control winding NC is wound in a direction perpendicular to each of the above windings. One end of the drive winding NB at the switching element Q1 side of the drive transformer PRT is connected to a capacitor C5 while the other end thereof is connected to an emitter of the switching element Q1. One end of the driving winding NB at the switching element Q2 side is connected to the ground while the other end thereof is connected to the capacitor C5, so that this driving winding NB is supplied with a voltage having the opposite polarity to that of the driving winding NB of the switching element Q1. One end of current detection winding ND is connected to a primary winding N3 of a magnetically-coupled transformer MCT, and the other end thereof is connected to a primary winding N1 of an insulating transformer PIT through a resonance capacitor CI.
PIT represents an insulating transformer for transmitting the switching outputs of the switching elements Q1 and Q2 to the secondary side, and one end of the primary winding N1 of the insulating transformer PIT is connected through the resonance capacitor C1 to the current driving winding ND in series while the other end thereof is grounded. An inductance component of the insulating transformer PIT which contains the resonance capacitor C1 and the primary winding N1 forms the resonance circuit. At the secondary side, the induced voltage which is induced at the secondary winding N2 by the output of the switching output flowing in the primary winding N1 is converted to a DC voltage by the bridge rectifying circuit D3 and the smoothing capacitor C3, and output as an output voltage Eo. A control circuit 1 compares the DC voltage output Eo of the secondary side with a reference voltage and supplies the DC current corresponding to the difference (comparison result) to the control winding NC of the drive transformer PRT as a control current.
In FIG. 2, MCT represents a magnetically-coupled transformer. In this magnetically-coupled transformer MCT, a secondary winding Ni (Li represents self-inductance) corresponding the choke coil CH in FIG. 1 and a winding N3 (inductance L3) corresponding to the tertiary winding of the insulating transformer PIT are set as a first winding, and closely coupled to each other, for example, in a winding ratio of 1:1 by a ferrite core. The primary winding N3 of the magnetically-coupled transformer MCT is connected to the primary winding N1 of the insulating transformer PIT in series through the resonance capacitor C1 and the current detection winding ND.
A switching operation of the switching power source thus constructed will be described.
First, upon power-on of the commercial alternating power source, base current is supplied to the bases of the switching elements Q1 and Q2 through the actuating resistors R6. If the switching element Q1 is previously switched on, the switching element Q2 is controlled to be switched off. At this time, resonance current flows from the current detection winding ND through the capacitor C1 to the primary winding N1 as the output of the switching element Q1. In this case, the switching elements Q2 and Q1 are controlled to be switched on and off respectively when the resonance current is equal to zero or a near value. At this time, the resonance current flows through the switching element Q2 in the opposite direction to the direction as described above. Subsequently, such a self-exciting switching operation that the switching elements Q1 and Q2 are alternately switched on is started, As described above, the switching elements Q1 and Q2 are repetitively alternately switched on and off using the terminal voltage of the smoothing capacitor as an operating power, whereby a drive current whose waveform is close to the resonance current waveform is supplied to the winding N1 of the primary side of the insulating transformer, and the alternating output is obtained at the winding N2 of the secondary side.
When the DC output voltage (Eo) at the secondary side is lowered, the current flowing in the control winding NC is controlled by the control circuit 1 so that the switching frequency is lowered (approaches the resonance frequency) and thus the drive current flowing in the primary winding N1 increases, thereby keeping the voltage constant.
As a power factor improving operation, the switching voltage corresponding to the resonance current flowing in the insulating transformer PIT is excited to a self-inductance Li of the secondary winding Ni by the primary winding N3 in the magnetically-coupled transformer MCT. Accordingly, the full-wave rectified voltage is superposed on the switching voltage at the winding Ni of the self-inductance Li, and then charged in a smoothing capacitor Ci. Accordingly, the terminal voltage of the smoothing capacitor Ci is reduced at the switching frequency by the amount of the superposed switching voltage. With this operation, the charge current flows for a period when the terminal voltage of the capacitor Ci is lower than the rectified voltage level of the bridge rectifying circuit. By setting the winding number of the magnetically-coupled transformer MCT or the like so that the period continues until it approaches a value near to zero volts, the power factor can be set approximately to 1. That is, the average alternating input current has a similar waveform to the AC voltage waveform, thereby improving the power factor.
In a power circuit using a magnetically-coupled transformer, the drive current of the insulating transformer PIT is reduced when a light load is imposed, and thus a small switching signal is induced at the secondary side of the magnetically-coupled transformer MC by the drive current. Accordingly, the level of the charging current is low when a light load is imposed, and it is high when a heavy load is imposed. Therefore, the terminal voltage of the smoothing capacitor can be prevented from abnormally increasing, particularly when the light load is imposed, and the improvement of regulation which is difficult using an ordinary MS system can be performed. Therefore, variation of the rectified smoothing voltage Vi due to variation of the alternating input voltage of VAC.+-.20% can be suppressed, so that it is unnecessary to aim at the improvement of breakdown voltages of the switching elements Q1 and Q2, the smoothing capacitor Ci, etc.
FIG. 3 is a circuit diagram showing a half bridge type switching power circuit of the self-exciting current resonance type, and the same elements as FIG. 2 are represented by the same reference numerals, and the description thereof is omitted.
In this circuit diagram, the drive transformer comprises a CDT (Converter Drive Transformer) for driving the switching elements Q1 and Q2 at a predetermined switching frequency. The converter transformer for transmitting the switching output of the primary side to the secondary side comprises an insulating transformer PRT (Power Regulation Transformer) which is provided with a control winding NC. The control winding NC is supplied with the control current corresponding to the DC voltage Eo from the control circuit 1 to change the saturated characteristic of the insulating transformer PRT and control leaking magnetic flux, thereby performing a constant-voltage control (it is also called as "series resonance frequency control system").
The primary winding N3 of the magnetically-coupled transformer MCT of this circuit is connected to both ends of the winding N4 at the secondary side of the insulating transformer PRT, so that it is supplied with a voltage having the switching frequency which is excited at the secondary winding N4. With this construction, the same improvement of the power factor as described with reference to FIG. 2 can be performed through the operation of the magnetically-coupled transformer MCT. The secondary winding N4 is connected to a full-wave rectifying smoothing circuit comprising diodes D4 and D5 and a capacitor C4, whereby a DC output voltage E1 can be supplied therefrom.
FIG. 4 is a perspective view showing the construction of the magnetically-coupled transformer MCT used in the embodiments of FIGS. 2 and 3. In this case, E-shaped cores CR1 and CR2 formed of ferrite material are combined with each other so that the magnetic legs thereof are confronted to each other, thereby forming an EE-shaped core. at this time, and a gap G is formed between the center magnetic legs of the E-shaped cores as shown in FIG. 4. A primary winding Ni and a secondary winding N3 are wound around the respective center magnetic legs to construct a magnetically-coupled transformer.
In the circuit shown in FIG. 2, the magnetically-coupled transformer needs some degree or size if it is to be endurable against a heavy load above 200 W, and thus it is difficult to miniaturize the magnetically-coupled transformer. Furthermore, in the circuit shown in FIG. 3, the primary winding Ni of the magnetically-coupled transformer MCT is a circuit element of the primary side of the insulating transformer PRT while the secondary winding N3 is connected to the low voltage output winding N4 of the secondary side of the insulating transformer PRT, so that it is required to perform a winding operation on the magnetically-coupled MCT while keeping an insulation distance between the primary winding Ni and the secondary winding N3. Therefore, miniaturization of the magnetically-coupled transformer is also difficult in this case.